DE102023211066A1 - MIMO radar system with novel phase modulation to suppress unwanted effects - Google Patents

Abstract

The present invention relates to a method for a radar system and a corresponding radar system for environmental detection, comprising transmitting means with parallel transmitting antennas for emitting transmitted signals containing one or more sequences of individual signals, means for changing the phase position of the transmitted individual signals, by means of which a phase change that differs for the transmitting antennas is realized across the individual signals, hereinafter referred to as phase modulation, wherein these phase modulation means can generate at least three different phase values, receiving means with one or more receiving antennas for receiving transmitted signals reflected from objects, and signal processing means for processing the received signals. The phase modulation is formed by the sum of a linear phase change and an irregular phase sequence, wherein the linear phase change has a different gradient across the transmitted signals of the transmitting antennas and serves to separate the components in the received signals caused by the transmitted signals of the different transmitting antennas, while the superimposed irregular component is identical across the transmitting antennas.

Description

The invention relates to a radar method for a radar system and a radar system for use in driver assistance systems in motor vehicles. The radar system has multiple transmit and receive antennas operated in parallel, which is referred to as MIMO (multiple-input multiple-output), and according to the invention uses a novel phase modulation (or the method according to the invention), which avoids undesirable effects, in particular ghost detections and falsification of measured values.

State of the art

Motor vehicles are increasingly being equipped with driver assistance systems that use sensor systems to detect the surroundings and, based on the detected traffic situation, derive automatic vehicle reactions and/or instruct the driver, particularly by issuing warnings. A distinction is made between comfort and safety functions.

FSRA (Full Speed Range Adaptive Cruise Control) plays an important role as a comfort feature in current development. The vehicle regulates its own speed to the desired speed set by the driver, provided the traffic situation permits; otherwise, the own speed is automatically adjusted to the traffic situation.

Safety functions now exist in a wide variety of forms. One group includes functions for reducing braking or stopping distances in emergency situations, all the way up to autonomous emergency braking. Another group is lane change functions: These warn the driver or intervene in the steering if the driver intends to make a dangerous lane change, i.e., if a vehicle in the adjacent lane is either in the blind spot (known as BSD - "Blind Spot Detection") or is rapidly approaching from behind (LCA - "Lane Change Assist").

Nowadays, however, the driver is no longer just assisted, but the driver's task is increasingly carried out autonomously by the vehicle, i.e. the driver is increasingly being replaced; this is referred to as autonomous driving.

Radar sensors are used for systems of the type described above, often in combination with sensors of other technologies, such as camera sensors. Radar sensors have the advantage, among other things, that they operate reliably even in poor weather conditions and can directly measure not only the distance of objects but also their radial relative velocity via the Doppler effect. Today, 77 GHz and 79 GHz are typically used as transmission frequencies.

The above-mentioned functions require high detection quality, which requires precise angle formation. Therefore, MIMO radars are increasingly being used. These radars feature multiple transmit and receive antennas operating in parallel, using all combinations of transmit and receive antennas to achieve the best possible angle formation. For parallel operation of the transmit antennas, their transmitted signals must be modulated differently so that the components of the received signals caused by them can be separated. The patent EP 2 629 113 B1 shows the approach of using a sequence of individual transmission signals and varying the phases of the individual transmission signals; as an example, a binary phase modulation is shown (i.e., with only two phase values, 0° and 180°), which either represents a random sequence or has a periodic, particularly alternating, course. It is also mentioned that the phase modulation can be composed of a deterministic and a random component. The disadvantage of using random phase modulations to differentiate between the transmission signals is that the integration of received signals across the sequence of individual transmission signals must be implemented separately for each transmitting antenna, which is a significant effort. This multiple integration avoids the linear phase modulation known in the prior art, in which the rate of change, i.e., the gradient of the linear phase change, varies across the transmitting antennas; however, this can lead to adverse effects due to inaccuracies in the phase modulation means and/or spectral convolutions.

Task, solution and advantages of the invention

The object of the invention is to provide an improved phase modulation for automotive MIMO radar systems, which at least largely suppresses the adverse effects occurring in current phase modulations.

This object is fundamentally achieved by a method according to claim 1 or a radar system according to claim 7. Expedient embodiments of the invention are claimed in the subclaims. The core idea is that a random phase modulation that is the same for all transmitting antennas is superimposed on the known linear phase modulation, whereby this random phase modulation uses more than two phase values (i.e., no binary phase modulation), thereby achieving numerous positive effects, in particular eliminating problems that would otherwise arise. The integration of received signals across the sequence of individual transmitted signals can be carried out jointly for all transmitting antennas, i.e., only once.

The advantages of the invention arise from the fact that the novel phase modulation ensures high detection quality and reduces hardware requirements, particularly with regard to the quality of the phase modulation means and the required digital computing power.

The radar system to which the inventive method for environmental detection relates includes 1) transmitting means with M _TX transmitting antennas operating in parallel for emitting transmitted signals which include one or more sequences of K individual signals whose general form is preferably the same or similar, 2) means for changing the phase position of the transmitted individual signals, by means of which a phase change which is different for the M _TX transmitting antennas is realized across the K individual signals, hereinafter referred to as phase modulation, wherein these phase modulation means can generate at least three different phase values which are preferably at least approximately evenly distributed over the phase unambiguousness range 2π, 3) receiving means with one or more receiving antennas for receiving transmitted signals reflected from objects, and 4) signal processing means for processing the received signals. The method is characterized in that the phase modulation is formed by the sum of a linear phase change, possibly apart from phase jumps due to the phase unambiguousness range of 2π, and an irregular phase sequence, wherein the linear phase change has a different slope across the transmission signals of the M _TX different transmitting antennas and serves to separate the components in the received signals caused by the transmission signals of the different transmitting antennas, while the superimposed irregular component is identical across the M _TX different transmitting antennas and serves in particular to reduce or prevent effects of inaccuracies in the phase modulation means and/or effects of spectral convolutions.

Conveniently, the superimposed irregular phase sequence can assume random or pseudorandom values.

Furthermore, in the digital signal processing means there can be an integration over K signals which are derived from the received signals to the K transmitted signals, whereby this integration is made only once for the signals to the M _TX different transmitting antennas and a compensation of the irregular phase component takes place beforehand.

An advantageous embodiment of the invention is characterized in that the integration over K signals, which are derived from the received signals to the K transmitted signals, is carried out as a discrete Fourier transformation, preferably in the form of a fast Fourier transformation, and the components corresponding to the various transmitting antennas lead to power peaks at different points of the resulting spectrum.

The individual transmission signals can expediently be linearly frequency modulated, with their center frequency possibly changing successively, or can represent OFDM signals or be generated with pseudorandom fast phase modulation.

Furthermore, the irregular portion of the phase modulation can also serve to decorrelate effects caused by radiation from receiving antennas due to limited isolation, effects of internal couplings and interference from other radar systems and thus reduce or prevent negative effects from them.

Brief description of the drawings

• 1 shows the exemplary embodiment of a radar system.
• In 2 the frequency modulation consisting of a sequence of frequency ramps is shown.
• 3a shows the magnitude of the two-dimensional spectrum in the object distance gate j ₀ , i.e. |Ŝ ₂ (jo,I,m _TX ,m _RX )|, without random phase modulation component, 3b with random phase modulation component; inaccuracies occurring in real phase shifters are assumed, i.e. small phase and amplitude errors.

Example

The exemplary design of a radar system is considered, which is 1 The radar system has M _TX = 3 transmitting antennas TX0-2 for transmitting transmitted signals and M _RX = 4 receiving antennas RX0-RX3 for receiving transmitted signals reflected from objects. The antennas are designed as patch antennas on a flat circuit board 1.1 using planar technology. This circuit board is oriented horizontally and vertically in the vehicle as shown in the image and faces the direction of travel. All antennas (transmitting and receiving antennas) have the same beam characteristics in elevation and azimuth. The M _RX = 4 receiving antennas (and thus their phase, i.e., radiation centers) each have the same lateral, i.e., horizontal, distance d = λ/2 = 1.96mm from each other, where λ = c/76.5GHz = 3.92mm is the mean wavelength of the radiated signals in the used frequency band of 76-77GHz and c = 3*10 ⁸ m/s is the speed of light. The M _TX = 3 transmitting antennas have four times the horizontal distance, i.e., 4d = 2λ, so that all combinations of transmitting and receiving antennas synthesize an equidistant array of M _TX ·M _RX = 12 antenna channels in a grid of d = λ/2, which is used to determine the azimuth angle of objects.

The transmission signals radiated by the transmitting antennas are obtained from the high-frequency oscillator 1.2 in the 76-77 GHz range, whose frequency can be varied via a control voltage v _control . The control voltage is generated in the control means 1.8, which control means contain, for example, a phase-locked loop or a digital-to-analog converter, which are controlled such that the frequency response of the oscillator corresponds to the desired frequency modulation. The phase position of the transmission signals can be individually adjusted and varied for the M _TX transmitting antennas via phase shifters 1.3; the phase shifters realize 64 at least approximately uniformly distributed phase values over the phase unambiguous range 2π. These phase shifters modulate the transmission signals from the different transmitting antennas differently, so that parallel transmission on all transmitting antennas, i.e. MIMO operation, is possible because after demodulation in the received signals the components originating from the different transmitting antennas can be separated.

The signals received by the four receiving antennas are also mixed down to the low frequency range in the real-valued mixers 1.4 with the signal from oscillator 1.2. The received signals then pass through the bandpass filters 1.5 with the transfer function shown, the amplifiers 1.6, and the analog/digital converters 1.7. They are then further processed in the digital signal processing unit 1.9.

In order to measure the distance of objects, as in 2 shown - the frequency f _TX of the high-frequency oscillator and thus of the transmitted signals changes linearly very quickly (in T _ch = 51.2 µs by B _ch = 600 MHz, with the center frequency f _c = 76.5 GHz); this is referred to as a frequency ramp (often also referred to as a "chirp"). The frequency ramps are repeated periodically in a fixed grid T _D = 70 µs; in total there are K = 512 frequency ramps, which all have the same frequency response, i.e. the same frequency gradient, the same frequency position (in particular the same start and center frequency) and the same duration. In recent years this type of modulation has become increasingly widespread and established in radar for detecting the environment of motor vehicles. It allows a high sensor range and speed resolution (through long data acquisition times) as well as a high distance resolution (through the use of high modulation bandwidth).

During each frequency ramp k=0,...,K-1, the received signals from each of the M _RX A/D converters are sampled I = 2048 times at intervals of T _s = 25ns (i.e. at 40MHz), whereby the sampling always begins at the same time relative to the start of the ramp (see 2 ); the digital samples resulting in the receive path m _RX with index i=0,..., I-1 are denoted by s(i,k,m _RX ). Signal sampling only makes sense in the time domain where received signals from objects arrive in the distance range of interest - after the ramp start, at least the propagation time corresponding to the maximum distance of interest must be waited for (for a maximum distance of interest of 200m, this corresponds to 1.33µs); it should be noted that here and in the following, distance always refers to the radial distance, and relative velocity to its radial component.

$${\text{mit j}}_0=\text{r/}\left(\text{Meter}\right)\cdot {\text{B}}_{\text{ch}}/150\text{Mhz}\text{,}$$

d. h. die Frequenz der Schwingung ist proportional zur Objektentfernung r (j₀ ist eine normierte Frequenz). Eine radiale Relativbewegung des Objekts zum Sensor bewirkt einen sich über die K = 512 Frequenzrampen ändernden Phasenlagenbeitrag φ_v(k) der sinusförmigen Schwingung; bei einer Bewegung mit konstanter radialer Geschwindigkeitskomponente v ergibt sich: $${\mathrm{\phi }}_{\text{v}}\left(\text{k}\right)=2\mathrm{\pi }\cdot \text{k/K}\cdot {\text{I}}_0{\text{ mit I}}_0=2{\text{KT}}_{\text{D}}{\text{vf}}_{\text{c}},$$

d. h., ein lineare Phasenlagenänderung über die Frequenzrampen k, wobei die Änderungsgeschwindigkeit der Phase proportional zur radialen Relativgeschwindigkeit v des Objekts ist. Der Phasenbeitrag φ_PM,TX(k,m_TX) in obiger Bez. (1) beschreibt die mit den Phasenschiebern 1.3 realisierte Phasenmodulation $${\mathrm{\phi }}_{\text{PM}\text{,TX}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)={\mathrm{\phi }}_{\text{PM}\text{,lin}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\text{PM}\text{,r}}\left(\text{k}\right),$$

welche zwei Anteile aufweist: Zum einen den sich linear über die Frequenzrampen k ändernden Anteil $${\mathrm{\phi }}_{\text{PM}\text{,lin}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)=2\mathrm{\pi }\cdot \text{k}\cdot \text{p}\left({\text{m}}_{\text{TX}}\right)/\text{P}$$

mit über die Sendepfade unterschiedlicher Änderungsgeschwindigkeit p(m_TX)/P mit ganzzahliger gemeinsamer Periode P und hier als ganzzahlig betrachteten normierten Änderungsgeschwindigkeiten p(m_TX) (normiert bezüglich langsamster Änderungsgeschwindigkeit 1/P); dieser Anteil dient zur späteren Trennung der von den verschiedenen Sendepfaden stammenden Anteile in den Empfangssignalen (siehe auch nachfolgende Erläuterungen). Zum anderen einen über die Frequenzrampen k zufälligen oder pseudozufälligen Anteil φ_PM,r(k), welcher für alle Sendepfade identisch ist und welcher durch eine für jede Frequenzrampe k zufällige Auswahl aus den 64 Phasenwerten der Phasenschieber 1.3 generiert wird; dieser erfindungsgemäße Anteil dient zur Unterdrückung unerwünschter Effekte, was später hergeleitet und erläutert wird. Die zwei hinteren, von den Frequenzrampen k unabhängigen Phasenanteile φ_α,TX(m_TX) und φ_α,RX(m_RX) stellen die vom Azimutwinkel α des Objekts abhängigen Phasenlagen für die verschiedenen Sende- und Empfangspfade dar. Es sei noch bemerkt, dass real realisierte Phasenwerte im Bereich 0...2π liegen, da sich wegen der zyklischen Eigenschaft von Phasen alle Phasenwerte in diesen Bereich abbilden lassen, was mathematisch gesehen eine Modulo-Funktionalität ist.As is known from the state of the art (see e.g. EP 2 629 113 B1 ) and can also be easily derived, the sampling signal ŝ(i,k,m _TX ,m _RX ) caused by the transmission signal of a transmitting antenna m _TX represents, in the case of a single point-like object at a distance r, a sinusoidal oscillation over the index i, which can be described in very good approximation as follows: $$\widehat{\text{s}}\left(\text{i}\text{,k}{\text{,m}}_{\text{TX}},{\text{m}}_{\text{RX}}\right)=\text{A}\cdot \text{sin}\left[2\mathrm{\pi }\cdot \text{i/}\underset{\_}{\text{l}}\cdot {\text{<figure-callout id="0" label="j" filenames="DE102023211066A1\_0012.png,DE102023211066A1\_0013.png" state="{{state}}">j</figure-callout>}}_0+{\mathrm{\phi }}_{\text{v}}\left(\text{k}\right)+{\mathrm{\phi }}_{\text{PM}\text{,TX}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{TX}}\left({\text{m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{RX}}\left({\text{m}}_{\text{RX}}\right)\right]$$

$${\text{mit j}}_0=\text{r/}\left(\text{Meter}\right)\cdot {\text{B}}_{\text{ch}}/150\text{Mhz}\text{,}$$

i.e., the frequency of the oscillation is proportional to the object distance r (j ₀ is a normalized frequency). A radial relative movement of the object to the sensor causes a phase position contribution φ _v (k) of the sinusoidal oscillation that changes over the K = 512 frequency ramps; for a movement with a constant radial velocity component v, the result is: $${\mathrm{\phi }}_{\text{v}}\left(\text{k}\right)=2\mathrm{\pi }\cdot \text{k/K}\cdot {\text{I}}_0{\text{ mit I}}_0=2{\text{KT}}_{\text{D}}{\text{vf}}_{\text{c}},$$

ie, a linear phase change over the frequency ramps k, where the rate of change of the phase is proportional to the radial relative velocity v of the object. The phase contribution φ _PM,TX (k,m _TX ) in the above equation (1) describes the phase modulation realized with the phase shifters 1.3 $${\mathrm{\phi }}_{\text{PM}\text{,TX}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)={\mathrm{\phi }}_{\text{PM}\text{,lin}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\text{PM}\text{,r}}\left(\text{k}\right),$$

which has two components: Firstly, the component which changes linearly over the frequency ramps k $${\mathrm{\phi }}_{\text{PM}\text{,lin}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)=2\mathrm{\pi }\cdot \text{k}\cdot \text{p}\left({\text{m}}_{\text{TX}}\right)/\text{P}$$

with different rates of change p(m _TX )/P across the transmission paths with an integer common period P and normalized rates of change p(m _TX ) considered here as integers (normalized with respect to the slowest rate of change 1/P); this component is used for the later separation of the components in the received signals originating from the various transmission paths (see also the following explanations). On the other hand, a random or pseudo-random component φ _PM,r (k) across the frequency ramps k, which is identical for all transmission paths and which is generated by a random selection for each frequency ramp k from the 64 phase values of the phase shifters 1.3; this component according to the invention is used to suppress undesired effects, which is derived and explained later. The two rear phase components φ _α,TX (m _TX ) and φ _α,RX (m _RX ), which are independent of the frequency ramps k, represent the phase positions for the various transmission and reception paths, which depend on the azimuth angle α of the object. It should also be noted that actually realized phase values lie in the range 0...2π, since due to the cyclic nature of phases, all phase values can be mapped into this range, which mathematically is a modulo functionality.

Finally, with regard to (1), it should be mentioned that the amplitude A of the received signal is assumed to be independent of the transmit and receive paths, i.e., all transmit and receive paths should be equally strong; this assumption has no influence on the further considerations.

wobei „exp“ die Exponentialfunktion bezeichnet und j die imaginäre Einheit ist, ergibt sich die DFT, also das Spektrum Š₁(j,k,m_TX,m_RX) des von einem Sendepfad bewirkten Abtastsignals ŝ(i,k,m_TX,m_RX) zu: $$\begin{array}{c}{\widehat{\text{S}}}_1\left(\text{j}\text{,k}{\text{,m}}_{\text{TX}},{\text{m}}_{\text{RX}}\right)=\text{A/}\left(2\widehat{\text{j}}\right).\\ [\text{exp}\left(\widehat{\text{j}}\left({\mathrm{\phi }}_{\text{v}}\left(\text{k}\right)+{\mathrm{\phi }}_{\text{PM}\text{,TX}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{TX}}({\text{m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{RX}}\left({\text{m}}_{\text{RX}}\right)\right))\cdot {\text{W}}_1\left({\text{mod}}_{\text{J}}\left(\text{j}-{\text{j}}_0\right)\right)\\ -\text{exp}\left(\mathrm{-}\widehat{\text{j}}({\mathrm{\phi }}_{\text{v}}\left(\text{k}\right)+{\mathrm{\phi }}_{\text{PM}\text{,TX}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{TX}}({\text{m}}_{\text{TX}}\right)\text{y}+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{RX}}\left({\text{m}}_{\text{RX}}\right)))\cdot {\text{W}}_1\left({\text{mod}}_{\text{J}}\left(\text{j}+{\text{j}}_0\right)\right)],\end{array}$$

wobei j=0,..., 1-1 die Laufvariable für den Bildbereich, also den Frequenzbereich der DFT ist und die sogenannten Entfernungstore repräsentiert (weil Frequenz des Empfangssignals proportional zu Entfernung ist), W₁(j) das Spektrum der verwendeten Fensterfunktion w₁(i) ist und „mod_J“ die Modulofunktion zum Modul J darstellt. Das Spektrum W₁(j) der Fensterfunktion hat eine recht scharfe Leistungsspitze bei j = 0, welche sich etwa über drei Frequenzwerte j erstreckt. Das von einem einzelnen Objekt bewirkte Spektrum Š₁(j,k,m_TX,m_RX) weist gemäß Bez. (6) also zwei Leistungsspitzen bei den Frequenzen j₀ und I-j₀ auf (unter der Annahme 0 ≤ j₀ ≤1, welche wegen nicht negativer Entfernungen und der Wirkung der Bandpassfilter 1.5 gültig ist). Wenn 0 ≤ j₀ ≤ 1/2 ist, dann trägt die Leistungsspitze bei I-j₀, also in der oberen Hälfte des Spektrums keine zusätzliche Information, was ganz allgemein für die obere Hälfte des Spektrums gilt, weil diese auf Grund des reellwertigen Eingangssignals konjugiert komplex gespiegelt zur unteren Hälfte ist; deshalb wird für die weitere Verarbeitung nur die untere Spektrumshälfte betrachtet, also nur die Frequenzen bzw. Entfernungstore j=0...I/2.In the digital signal processing unit 1.9, a first discrete Fourier transform (DFT) is performed for the received signals s(i,k,m _RX ) per frequency ramp k and receive path m _RX after multiplication with a suitable window function w ₁ (i) over the time index i=0,..., I-1, since this corresponds to optimal filtering for the signal form (1); the DFT is expediently implemented with a fast Fourier transform (FFT). With the relationship $$\text{sin}\left(\text{x}\right)=\left(\text{exp}\left(\widehat{\text{j}}\cdot \text{x}\right)-\text{exp}\left(\widehat{\text{j}}\cdot \text{x}\right)\right)/\left(2\widehat{\text{j}}\right),$$

where “exp” denotes the exponential function and j is the imaginary unit, the DFT, i.e. the spectrum Š ₁ (j,k,m _TX ,m _RX ) of the sampling signal ŝ(i,k,m _TX ,m _RX ) produced by a transmission path, is: $$\begin{array}{c}{\widehat{\text{S}}}_1\left(\text{j}\text{,k}{\text{,m}}_{\text{TX}},{\text{m}}_{\text{RX}}\right)=\text{A/}\left(2\widehat{\text{j}}\right).\\ [\text{exp}\left(\widehat{\text{j}}\left({\mathrm{\phi }}_{\text{v}}\left(\text{k}\right)+{\mathrm{\phi }}_{\text{PM}\text{,TX}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{TX}}({\text{m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{RX}}\left({\text{m}}_{\text{RX}}\right)\right))\cdot {\text{W}}_1\left({\text{mod}}_{\text{J}}\left(\text{j}-{\text{j}}_0\right)\right)\\ -\text{exp}\left(\mathrm{-}\widehat{\text{j}}({\mathrm{\phi }}_{\text{v}}\left(\text{k}\right)+{\mathrm{\phi }}_{\text{PM}\text{,TX}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{TX}}({\text{m}}_{\text{TX}}\right)\text{y}+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{RX}}\left({\text{m}}_{\text{RX}}\right)))\cdot {\text{W}}_1\left({\text{mod}}_{\text{J}}\left(\text{j}+{\text{j}}_0\right)\right)],\end{array}$$

where j=0,..., 1-1 is the control variable for the image domain, i.e. the frequency domain of the DFT and represents the so-called distance gates (because the frequency of the received signal is proportional to the distance), W ₁ (j) is the spectrum of the window function w ₁ (i) used and “mod _J ” represents the modulo function for the modulo J. The spectrum W ₁ (j) of the window function has a very sharp power peak at j = 0, which extends over approximately three frequency values j. The spectrum Š ₁ (j,k,m _TX ,m _RX ) caused by a single object therefore has, according to equation (6), two power peaks at the frequencies j ₀ and Ij ₀ (assuming 0 ≤ j ₀ ≤1, which is valid due to non-negative distances and the effect of the bandpass filters 1.5). If 0 ≤ j ₀ ≤ 1/2, then the power peak at Ij ₀ , i.e. in the upper half of the spectrum, carries no additional information, which generally applies to the upper half of the spectrum because, due to the real-valued input signal, it is a complex conjugate mirror image of the lower half; therefore, only the lower half of the spectrum is considered for further processing, i.e. only the frequencies or distance gates j=0...I/2.

Before a second DFT can be performed over dimension k, the random phase component φ _PM,r (k) of the phase modulation φ _PM,TX (k,m _TX ) must be compensated according to Equation (4a) (the remaining phase components represent a linear phase progression over dimension k, which is necessary for the application of the DFT as optimal filtering). For the range of interest j ₀ = 0...I/2 (i.e., objects located there), the first term in Equation (6) is relevant (only the lower half of the spectrum j = 0...I/2 is considered), so that to compensate for the random phase modulation component, it must be multiplied by exp(-ĵ φ _PM,r (k)); this gives: $$\begin{array}{l}{\widehat{\text{S}}}_{\mathrm{1,}\text{komp}}\left(\text{j}\text{,k}{\text{,m}}_{\text{TX}},{\text{m}}_{\text{RX}}\right)=\text{A/}\left(2\widehat{\text{j}}\right).\\ [\text{exp}\left(\widehat{\text{j}}\left({\mathrm{\phi }}_{\text{v}}\left(\text{k}\right)+{\mathrm{\phi }}_{\text{PM}\text{,lin}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{TX}}({\text{m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{RX}}\left({\text{m}}_{\text{RX}}\right)\right))\cdot {\text{W}}_1\left({\text{mod}}_{\text{J}}\left(\text{j}-{\text{j}}_0\right)\right)\\ \\ -\text{exp}(\mathrm{-}\widehat{\text{j}}({\mathrm{\phi }}_{\text{v}}\left(\text{k}\right)+{\mathrm{\phi }}_{\text{PM}\text{,lin}}\left(\text{k}{\text{,m}}_{\text{TX}}\right)+2{\mathrm{\phi }}_{\text{PM}\text{,r}}\left(\text{k}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{TX}}\left({\text{m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{RX}}({\text{m}}_{\text{RX}})))\cdot {\text{W}}_1\left({\text{mod}}_{\text{J}}\left(\text{j}+{\text{j}}_0\right)\right)\\ ].\end{array}$$

wobei I=0,..., K-1 die Laufvariable für den Bildbereich, also den Frequenzbereich der zweiten DFT ist und die sogenannten Dopplertore repräsentiert (weil Frequenz proportional zu Relativgeschwindigkeit ist, abgesehen vom zusätzlichen Phasenmodulationsanteil p(m_TX)·K/P), W₁₂(j,I) das zweidimensionale Spektrum der verwendeten zweidimensionalen Fensterfunktion w₁(i)·w₂(k) ist (hat Leistungsspitze bei j = 0 und I = 0) und R_PM(I) das Spektrum des einen zufälligen Phasenverlauf aufweisenden Einheitszeigers exp(-ĵ2φ_PM,r(k)) darstellt (2φ_PM,r(k) nimmt in zufälliger Weise Werte aus 32 verschiedenen über 2π gleichverteilten Phasenwerten an - dabei ist schon die Moduloeigenschaft der Phase berücksichtigt); damit ist auch R_PM(I) selber ein Rauschen, welches Rayleigh-verteilt ist. Für ein Objekt im interessierenden Entfernungstorbereich j₀=0...I/2 ergibt sich im zweidimensionalen Spektrum Ŝ₂(j,I,m_TX,m_RX) nach Bez. (8) vom ersten Term eine Leistungsspitze beim Entfernungstor j = j₀ und Dopplertor I = I₀+p(m_TX)·K/P; der zweite Term erzeugt bei j = I-j₀ (und seiner unmittelbaren Nachbarschaft) ein über alle Dopplertore k verteiltes Rauschen, wobei dieses j außerhalb dem interessierenden und betrachteten Entfernungstorbereich j=0...I/2 liegt.For each distance gate j and receive path m _RX and after multiplication with a window function w ₂ (k), a second DFT is then carried out, this time over the frequency ramp index k (preferably again over an FFT); this results in the two-dimensional spectrum $$\begin{array}{c}{\widehat{\text{S}}}_2\left(\text{j}\text{,l}{\text{,m}}_{\text{TX}},{\text{m}}_{\text{RX}}\right)=\text{A/}\left(2\widehat{\text{j}}\right).\\ [\text{exp}\left(\widehat{\text{j}}\left({\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{TX}}\left({\text{m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{RX}}({\text{m}}_{\text{RX}}\right)\right))\cdot {\text{W}}_{12}({\text{mod}}_{\text{J}}\left(\text{j}-{\text{j}}_0\right),{\text{mod}}_{\text{K}}\left(\text{I}-{\text{I}}_0-\text{p}\left({\text{m}}_{\text{TX}}\cdot \text{K/P}\right)\right)\\ -{\text{R}}_{\text{PM}}\left(\text{I}+{\text{I}}_0+\text{p}\left({\text{m}}_{\text{TX}}\right)\cdot \text{K/P}\right)\cdot \text{exp}\left(-\widehat{\text{j}}\left({\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{TX}}\left({\text{m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{RX}}({\text{m}}_{\text{RX}}\right)\right))\cdot {\text{W}}_1\left({\text{mod}}_{\text{J}}\left(\text{j}+{\text{j}}_0\right)\right)],\end{array}$$

where I=0,..., K-1 is the control variable for the image domain, i.e. the frequency domain of the second DFT and represents the so-called Doppler gates (because frequency is proportional to relative speed, apart from the additional phase modulation component p(m _TX )·K/P), W ₁₂ (j,I) is the two-dimensional spectrum of the used two-dimensional window function w ₁ (i)·w ₂ (k) (has a power peak at j = 0 and I = 0) and R _PM (I) represents the spectrum of the unit vector exp(-ĵ2φ _PM,r (k)) which has a random phase curve (2φ _PM,r (k) randomly takes on values from 32 different phase values evenly distributed over 2π - the modulo property of the phase is already taken into account); thus R _PM (I) itself is noise which is Rayleigh distributed. For an object in the range of interest j ₀ =0...I/2, the two-dimensional spectrum Ŝ ₂ (j,I,m _TX ,m _RX ) according to equation (8) results from the first term a power peak at the range gate j = j ₀ and Doppler gate I = I ₀ +p(m _TX )·K/P; the second term produces at j = Ij ₀ (and its immediate divisible neighborhood) a noise distributed over all Doppler gates k, where this j lies outside the interesting and considered distance gate range j=0...I/2.

Now, an object is considered above the range gate of interest, i.e., in the range j ₀ = (I/2+1)...(I-1); because of the comparatively low attenuation in the transition region of the bandpass filters 1.5, objects can be detected particularly slightly above j ₀ = I/2 - this is referred to as overrange. Then, in the two-dimensional spectrum Ŝ ₂ (j,I,m _TX ,m _RX ) according to equation (8), the first term results in a power peak at j = j ₀ above the considered range gate j=0...I/2, while the second term, at the range gate j = Ij ₀ in the range gate of interest and considered j=0...I/2, generates noise distributed across all Doppler gates k. However, this noise will not lead to detection even if it is significantly higher than the system noise, since detections are only made for power peaks in dimension k that are significantly higher than the overall noise level there. If the random phase modulation component φ _PM,r (k) were not present, the two-dimensional spectrum Ŝ ₂ (j,I,m _TX ,m _RX ) would be as follows (can be seen from equations (7) and (8)): $$\begin{array}{c}{\widehat{\text{S}}}_2\left(\text{j}\text{,l}{\text{,m}}_{\text{TX}},{\text{m}}_{\text{RX}}\right)=\text{A/}\left(2\widehat{\text{j}}\right).\\ [\text{exp}\left(\widehat{\text{j}}\left({\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{TX}}\left({\text{m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{RX}}({\text{m}}_{\text{RX}}\right)\right))\cdot {\text{W}}_{12}({\text{mod}}_{\text{J}}\left(\text{j}-{\text{j}}_0\right),{\text{mod}}_{\text{K}}\left(\text{I}-{\text{I}}_0-\text{p}\left({\text{m}}_{\text{TX}}\cdot \text{K/P}\right)\right)\\ -\text{ex}\text{p}\left(-\widehat{\text{j}}\left({\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{TX}}\left({\text{m}}_{\text{TX}}\right)+{\mathrm{\phi }}_{\mathrm{\alpha }\mathrm{,}\text{RX}}({\text{m}}_{\text{RX}}\right)\right))\cdot {\text{W}}_{12}({\text{mod}}_{\text{J}}\left(\text{j}+{\text{j}}_0\right),{\text{mod}}_{\text{K}}\left(\text{I}-{\text{I}}_0-\text{p}\left({\text{m}}_{\text{TX}})\cdot \text{K/P}\right)\right)].\end{array}$$

Thus, an overrange, i.e. an object in the range j ₀ =(I/2+1)...(I-1), would now lead to a power peak at the distance gate j = Ij ₀ in the interesting and considered distance gate range j=0...I/2 through the second term; thus, a detection would be erroneously formed which not only has an incorrect (too short) distance, but also whose measured values for relative velocity and angle are incorrect (due to incorrect signs of the contributing quantities in the second term of ref. (9)).

The random phase modulation component φ _PM,r (k) thus prevents false detections due to overreaching; the noise generated instead is approximately 26 dB lower in terms of its average power than the peak power that would occur without the random component (the 26 dB results from the DFT integration gain of 10 log ₁₀ (K = 512) = 27 dB minus approximately 1 dB window loss). It should also be noted that the above considerations do not apply to a binary phase shifter, i.e., one with only the two states 0 and π; This is because the phase component 2φ _PM,r (k) in the second term of Equation (7) is always 0 (phase 2π corresponds to phase 0) and thus has no effect. Therefore, the second term of the two-dimensional spectrum Ŝ ₂ (j,I,m _TX ,m _RX ) does not represent noise, but rather, according to Equation (9), also generates a power peak, which then leads to the undesirable overreach effect. Therefore, at least three different phase values are required to convert overreach into noise.

Overreach problems arise from a folding effect, i.e. the mapping of frequencies onto other frequencies. If frequencies at other frequencies still have certain power components, this is referred to as overfolding - collectively, such effects are also referred to as spectral folding effects. Such an overfolding effect occurs with very close objects; as an example, consider an object at a distance gate j ₀ = 0.5 (it should be noted that signals from such close objects can be received despite the high attenuation of the 1.5 bandpass filters, since they generate a very strong signal at the receiving antenna due to the short distance). Without the random phase modulation component φ _PM,r (k), components of both terms would then be effective in the two-dimensional spectrum Ŝ ₂ (j,I,m _TX ,m _RX ) according to Equation (9) at a distance gate j = 0 (the power peak with the shape of the two-dimensional window spectrum W ₁₂ has a certain width and typically extends over 3 distance gates); the first term provides the correct information, while the second term represents false information. These erroneously occurring components of the second term can either lead to falsification of the measured values of the object (if they overlap with the real components of the first term and then, for example, influence the distance interpolation) or to non-real detections, i.e., ghost detections (if they do not overlap with the real components of the first term and thus form independent power peaks). Due to the random phase modulation component φ _PM,r (k), the second term of the two-dimensional spectrum Ŝ ₂ (j,I,m _TX ,m _RX ) represents noise according to Equation (8), so that no ghost detections occur; and because the noise is far below the real power peaks of the first term, the influence on the measured values of the object (distance, relative speed, and angle) is negligible.

Negative effects of spectral convolutions are thus avoided by the random phase modulation component φ _PM,r (k).

A fundamental problem with phase modulation is that the means used for it, especially phase shifters, are never ideal and therefore always contain certain errors. The 64 phase values of the phase shifters considered here, which are ideally evenly distributed over the phase range 2π = 360°, should have a standard deviation of 5°; and in addition, the amplitude of the realized vectors should also have a standard deviation of 10%. The modulation period used should be P = 4, and for the transmission path m _TX under consideration, the modulation rate of change should be p(m _TX ) = 1. Without a random modulation component, the four target phase values 0, 90°, 180°, and 270°, which are generated with the phase shifter indices 0, 16, 32, and 48, must then be repeated periodically according to equation (4b); These should have the real phase values -5°, 91°, 185° and 263° and the amplitude values 1.02, 1.1, 0.85 and 1.11. For an object at a distance gate j ₀ < I/2 and at a Doppler gate I ₀ = 72, the resulting two-dimensional spectrum in the object distance gate j ₀ , i.e. Š ₂ (j ₀ ,I,m _TX ,m _RX ), is 3a represented in magnitude and in dB; in addition to the regular power peak at I = I ₀ +K/P = 200, further power peaks occur in the grid K/P = 128 (harmonic frequencies of period P = 4), which are generated by the non-ideal phase shifter values and their periodic repetition and can lead to false detections, i.e., ghost detections. If the harmonics coincide with the correct power peaks of other transmitting antennas, they can falsify the angle formation result. It should also be mentioned that in 3a the noise lying well below the power peaks comes from system noise, which is not represented in the formulas above.

When using the inventive random superimposed phase component φ _PM,r (k) of the phase modulation φ _PM,TX (k,m _TX ) according to equation (4a), four phase values are no longer periodically repeated, but quasi-random phase values are used (the linear phase component is no longer visible); thus, the phase modulation errors (phase and amplitude) are no longer periodic, but quasi-random, so that in the spectrum Ŝ ₂ (j ₀ ,I,m _TX ,m _RX ) according to 3b No more harmonic power peaks occur – their energy is distributed in noise. The superimposed random phase modulation component φ _PM,r (k) thus avoids ghost detections caused by the inaccuracies that always occur in the phase modulation means and thus allows the use of relatively poor phase shifters, which can lead to cost reduction.

In addition to the previously described advantages of the random phase modulation component φ _PM,r (k), this also means that radiation via the receiving antennas (due to their limited TX isolation), internal couplings between the transmit and receive paths as well as interference from other radar systems are decorrelated and thus converted into noise - they cannot therefore significantly degrade the measurement quality of objects and cannot generate ghost detections.

So far, only the contribution from one transmit path m _TX has been considered. The two-dimensional total spectrum S ₂ (j,I,m _RX ) generated by all M _TX = 3 transmit paths is the sum of the individual contributions represented in the above formulas (i.e., the sum over m _TX = 0,...,M _TX -1). Because the random phase modulation component φ _PM,r (k) is constant across all M _TX = 3 transmit paths, its compensation and the second DFT need to be calculated only once for all transmit paths, not separately for each one (the latter would be necessary if the random phase modulation component differed between the transmit paths, which would result in significantly increased computational effort). The modulation used to differentiate the transmit paths—i.e., the linear phase change φ _PM,lin (k,m _TX ) that differs between the transmit paths—also does not require multiple second DFTs, as it does not need to be compensated prior to the DFT. Without compensation in the DFT, it simply leads to a corresponding shift in the power peaks. After this second DFT, which is performed jointly for all transmit paths (i.e., only once), M _TX power peaks result at the positions corresponding to the transmit paths (i.e., their linear phase change). This allows the separation of the components caused by the different transmit paths, which is necessary for MIMO operation (i.e., the use of all combinations of transmit and receive antennas for angle formation when the transmit antennas are operated in parallel). The advantages and effects of the random phase modulation component φ _PM,r (k) presented above are of course also retained in the overall spectrum of all M _TX = 3 transmission paths (they apply to each individual transmission path and thus also to their sum).

So far, only a single point-like object has been considered. The above considerations remain valid even in the case of multiple and/or extended objects (since this only represents a linear superposition of several individual signals).

The radar system considered so far has several receiving antennas and thus several receiving paths; of course, only one receiving antenna would also be possible, but this is generally not very advantageous (poor angle formation and unfavorable, i.e., unbalanced utilization of the hardware resources used, since transmission paths are complex to implement).

So far, a real-valued mixer has been considered. In a complex-valued mixer (also called an IQ mixer), there is ideally only a single power peak; however, in reality, the IQ generation is not entirely perfect, so there is also a smaller power peak at negative frequency, which can lead to ghost detections or corrupted measured values. The random phase modulation component φ _PM,r (k) converts this power peak into noise, thus avoiding any negative effects.

In the modulation considered so far according to 2 all frequency ramps have the same frequency position, i.e. start frequency (and thus center frequency). For improved distance resolution, as shown in DE 10 2020 210 079 B3 As described above, the starting frequency can be changed linearly via the frequency ramps, with the spacing of the frequency ramps preferably also changing linearly. This has no effect on the phase modulation according to the invention and its advantages; they therefore remain unchanged.

So far, a linear frequency ramp has been considered a sequentially repeated transmit signal. Instead of a frequency ramp, however, other signal forms can also be used, for example, a signal with pseudorandom binary phase modulation (i.e., the sign changes very quickly and pseudorandomly within the signal) or an OFDM signal (OFDM = Orthogonal Frequency Division Multiplexing). The sample values (after analog-to-digital conversion) cannot be used directly as input values for the first DFT above; instead, the Fourier transform must first be calculated for each transmit signal and receive antenna and divided by the spectrum of the transmit signal.

Finally, it should be noted that it is obvious to a person skilled in the art how the inventive considerations and embodiments presented in the above application example can be transferred to general measurements and parameter interpretations, i.e. they can also be applied to other numerical values.

QUOTES CONTAINED IN THE DESCRIPTION

This list of documents submitted by the applicant was generated automatically and is included solely for the convenience of the reader. This list is not part of the German patent or utility model application. The DPMA assumes no liability for any errors or omissions.

Cited patent literature

• EP 2 629 113 B1 [0007, 0022]
• DE 10 2020 210 079 B3 [0039]

Claims (7)

Method for a radar system for environmental detection, comprising - transmitting means with M _TX transmitting antennas operating in parallel for emitting transmitted signals which contain one or more sequences of K individual signals whose general form is preferably the same or similar, - means for changing the phase position of the transmitted individual signals, by means of which a phase change which is different for the M _TX transmitting antennas is realized across the K individual signals, hereinafter referred to as phase modulation, wherein these phase modulation means can generate at least three different phase values which are preferably at least approximately evenly distributed across the phase unambiguousness range 2π, - receiving means with one or more receiving antennas for receiving transmitted signals reflected by objects, - and signal processing means for processing the received signals, characterized in that the phase modulation is formed by a sum of a linear phase change, possibly apart from phase jumps due to the phase unambiguousness range of 2π, and an irregular _phase sequence, wherein the linear phase change has a different slope and serves to separate the components in the received signals caused by the transmission signals of the different transmitting antennas, while the superimposed irregular component is identical across the M _TX different transmitting antennas and serves in particular to reduce or prevent effects of inaccuracies in the phase modulation means and/or effects of spectral convolutions. Procedure according to Claim 1 , in which the superimposed irregular phase sequence assumes random or pseudorandom values. Method according to one of the above claims, in which an integration is carried out in the digital signal processing means over K signals which are derived from the received signals to the K transmitted signals, this integration being carried out only once for the signals to the M _TX different transmitting antennas and a compensation of the irregular phase component taking place beforehand. Procedure according to Claim 3 , in which the integration over K signals, which are derived from the received signals to the K transmitted signals, is carried out as a discrete Fourier transformation, preferably in the form of a fast Fourier transformation, and the components corresponding to the different transmitting antennas lead to power peaks at different points in the resulting spectrum. Method according to one of the above claims, in which the individual transmission signals are linearly frequency modulated, with their center frequency possibly changing successively, or represent OFDM signals or are generated with pseudorandom fast phase modulation. Method according to one of the above claims, in which the irregular portion of the phase modulation also serves to decorrelate effects due to radiation via receiving antennas caused by limited isolation, effects of internal couplings and interference from other radar systems and thus to reduce or prevent negative effects caused by them. Radar system for environmental detection, comprising - transmitting means with M _TX transmitting antennas operating in parallel for emitting transmitted signals which contain one or more sequences of K individual signals whose general form is preferably the same or similar, - means for changing the phase position of the transmitted individual signals, by means of which a phase change which is different for the M _TX transmitting antennas is realized across the K individual signals, hereinafter referred to as phase modulation, wherein these phase modulation means can generate at least three different phase values which are preferably at least approximately evenly distributed over the phase unambiguousness range 2π, - receiving means with one or more receiving antennas for receiving transmitted signals reflected by objects, - and signal processing means for processing the received signals, characterized in that the phase modulation is achieved by a sum of a linear phase change, possibly apart from phase jumps due to the phase unambiguousness range of 2π, and an irregular phase sequence is formed, wherein the linear phase change has a different slope across the transmission signals of the M _TX different transmitting antennas and serves to separate the components in the received signals caused by the transmission signals of the different transmitting antennas, while the superimposed irregular component across the M _TX different transmitting antennas is identical and in particular serves to reduce or prevent effects of inaccuracies in the phase modulation means and/or effects of spectral convolutions.

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