CN102088875A - A flow sensor system - Google Patents

Abstract

There is provided a flow sensor system for sensing fluid flow indicative of a puff in an aerosol generating system. The sensor system includes a sensing circuit comprising a sensing resistor and a voltage output. The sensing resistor is arranged to detect fluid flow based on a change in resistance. The sensing circuit is arranged such that the change in resistance of the sensing resistor causes a change in the voltage output. The sensor system also includes a signal generator arranged to supply a pulsed driving signal to the sensing circuit for powering the sensing circuit. The sensing circuit is powered when the pulsed driving signal is high and not powered when the pulsed driving signal is low. The sensor system is arranged to operate in a first mode, in which no puff is expected or detected and in which the pulsed driving signal has a first frequency, and a second mode, in which a puff is expected or detected and in which the pulsed driving signal has a second frequency, greater than the first frequency.

Description

Flow Sensor System

The invention relates to a flow sensor system. In particular, but not exclusively, the invention relates to a flow sensor for an aerosol generating system. The invention is particularly useful as a flow sensor system for smoking systems, such as electrically heated smoking systems.

For example US-A-5060671, US-A-5388594, US-A-5505214, US-A-5591368, WO-A-2004/043175, EP-A-0358002, EP-A-0295122, EP-A-1618803 A number of prior art documents, EP-A-1736065 and WO-A-2007/131449, disclose electrically operated smoking systems with a number of advantages. One advantage is that it significantly reduces sidestream smoke while allowing the smoker to selectively discontinue and resume smoking.

Prior art aerosol generating systems may include an aerosol-forming substrate, one or more heating elements for heating the substrate to form aerosol, and a power source to power the one or more heating elements. Prior art aerosol generating systems may provide a pulse of energy to the heater to provide a desired temperature range of operation and release a volatile compound for each puff. Many prior art aerosol generating systems include flow sensors for sensing fluid flow, such as airflow or smoke flow, in the aerosol generating system. The sensor could play an important role in managing smoke delivery. When the flow sensor senses an airflow indicative of an inhalation induced by a puff by the user, an aerosolization mechanism, which may include a heating element or any type of nebulizer, is activated to provide aerosol for that puff. The flow sensor may be a passive (ie, mechanical) sensor or an active sensor.

Passive sensors typically include a displacing membrane and electrical contacts. The airflow generated by the user's inhalation displaces the membrane so that it touches the electrical contacts, which activates the aerosolization mechanism. As long as the air flow is strong enough to keep the membrane displaced, the aerosolization mechanism will remain activated. The advantages of passive sensors include design simplicity, resulting low cost, and negligible power consumption. Active sensors are often based on heat losses caused by fluid flow. Such active sensors are often referred to as thermal anemometers. The sensor includes a resistor that is heated to a high temperature. As the flow cools the resistor, the resulting drop in temperature for a given power, or increase in power to maintain a given temperature, dictates the airflow velocity. The resistors are typically silicon microelectromechanical systems (MEMS) based resistors. The advantages of an active sensor include the fact that the heat loss is proportional to the flow rate, so this sensor can be used to provide information about the suction characteristics. In addition, the sensor is less affected by mechanical shocks during transport and handling.

Since flow sensors provided in prior art aerosol generating systems, including those described above, do suffer from a number of disadvantages, it is an object of the present invention to provide an improved flow sensor system suitable for aerosol generating systems.

According to a first aspect of the present invention there is provided a flow sensor system for sensing fluid flow indicative of a draw in an aerosol generating system, the sensor system being arranged to operate in a first mode and in a second mode, in which In a first mode, a puff is not expected or detected, and in a second mode, a puff is expected or detected, and includes a sensing circuit that includes a sense resistor and a voltage output, the sense resistor arranged to detect fluid flow based on a change in resistance, the sensing circuit being arranged such that a change in resistance of the sense resistor causes a change in the voltage output; and a signal generator arranged to provide the The sensing circuit provides a pulsed driving signal _S1 to power the sensing circuit such that the sensing circuit is powered when the pulsed driving signal _S1 is high and is not powered when the pulsed driving signal _S1 is low, wherein , the pulse driving signal S ₁ has a first frequency f ₁ in the first mode, and has a second frequency f ₂ greater than the first frequency f ₁ in the second mode.

Since the sensor system includes a sensing resistor incorporated into the sensing circuit, which has an output voltage as a difference voltage, the sensitivity is high and small flow changes can be detected. The use of a pulsed drive signal S ₁ means that the sensing circuit is not powered constantly, but only when the pulsed drive signal S ₁ is high, ie when the square wave signal S ₁ is 1 and not 0. This significantly reduces power consumption. The sensor system can be constantly active, which means no separate on/off switch is required. Frequencies _f1 and _f2 can be chosen to provide appropriate sensitivity and power consumption. A sensor system can be used to obtain qualitative and quantitative information about the puff.

The signal generator for providing the pulsed drive signal preferably comprises a microcontroller, the pulsed signal being provided on one output of the microcontroller. If the signal generator includes a microcontroller, the microcontroller is preferably programmed to control the values of _f1 and _f2 . In other embodiments, the signal generator used to provide the pulsed drive signal may be any type of programmable electronic circuit.

Preferably, the flow sensor system further comprises a current source arranged to provide a predetermined value of current through the sensing circuit, wherein the pulsed drive signal _S1 is provided to the current source. The predetermined value current source allows the use of the sense resistor in the sense circuit at a constant current, which provides a method of operation with minimal power consumption. Since the current source is powered via the pulsed drive signal _S1 , the current source is not powered constantly, but only when the pulsed signal is high, which further reduces power consumption. The current source reduces the non-linearity of the dependence of the voltage output of the sense circuit on the resistance of the sense resistor. In a preferred embodiment, the current source is a temperature compensated current source. This is advantageous because this eliminates any change in the voltage output of the sensing circuit if the ambient temperature changes. In one embodiment, the current source includes a voltage source, two transistors in a mirror construction, and an input resistor.

Preferably, the flow sensor further comprises a differential amplifier arranged to amplify the voltage output of the sensing circuit. This is advantageous since the output of the sensing circuit may only be a few millivolts. The differential amplifier preferably has low power consumption and high gain.

Preferably, the differential amplifier can be disabled when the pulse driving signal S ₁ is low and can be enabled when the pulse driving signal S ₁ is high. This further reduces power consumption. Preferably, the differential amplifier output is proportional to the voltage output of the sensing circuit within a certain range of voltage output values of the sensing circuit, and saturates when the voltage output of the sensing circuit is less than or greater than the range. That is, when the voltage output of the sensing circuit is less than the range, the differential amplifier output has a constant value; when the voltage output of the sensing circuit is greater than the range, the differential amplifier output has a constant value; and when the sensing circuit's voltage output When the voltage output is within this range, there is a linear relationship between the output of the sensing circuit and the output of the differential amplifier.

Preferably, the sensor system operates in the second mode for a predetermined period of time after a change in the voltage output of the sensing circuit indicative of a puff has been detected, and in the first mode at all other times. Thus, when a puff is detected or at another time, the pulsed drive signal S ₁ changes from the first frequency f ₁ to a second, higher frequency f ₂ . This means that when the sensor is operating in the first mode, the maximum time to puff is 1/f ₁ second. _f1 may be chosen to provide an appropriate balance between power consumption and sensitivity in the first mode. If a puff is detected while the sensor is operating in the second mode, the maximum time to puff is 1/f ₂ seconds. _f2 may be chosen to provide an appropriate balance between power consumption and sensitivity in the second mode. In one embodiment, the first frequency f ₁ is 3 Hz and the second frequency f ₂ is 22 Hz.

Preferably, the predetermined period of time for which the sensor operates in the second mode after a puff is detected is equal to the average time between puffs for a particular user. Additionally, the predetermined time may be adaptive such that it is continuously adjusted based on a moving average of previous times between puffs. Alternatively, the predetermined period of time may have a fixed value.

If the means for providing the pulsed drive signal _S1 comprises a microcontroller, preferably the voltage output of the sensing circuit is provided to an input of the microcontroller. This can be achieved via a differential amplifier. Then, in one embodiment, when the input to the microcontroller indicates that a puff has been detected, the microcontroller can change the pulsed drive signal _S1 on its output from the first frequency _f1 to the second frequency _f2 .

Preferably, other components in the aerosol generating system are provided with _a signal _S2 which is high when the voltage output of the sensing circuit indicates puff detection and which is high when the _voltage output of the sensing circuit indicates no puff is detected. It's low when it comes to puffing. If the means for providing the pulsed drive signal _S1 comprises a microcontroller, preferably the signal _S2 is provided on another output of the microcontroller. Preferably, the voltage output of the sensing circuit is provided to the input of the microcontroller. Then, when the input to the microcontroller indicates that a puff is detected, the microcontroller is set to output a high signal _S2 , and when the input to the microcontroller indicates that a puff is not detected, the microcontroller is set to output Low signal S ₂ . Other components in the aerosol generating system may include, but are not limited to, an aerosolization mechanism (which may be a vaporization mechanism, vaporization engine, atomization mechanism, or atomization engine), a nebulizer, a heating element, and a puff indicator.

The flow sensor system may also include means for adjusting the sensitivity of the sensor system comprising one or more of: a variable resistor in the sensing circuit; a self-adjusting offset circuit ; and a signal generator for providing a pulsed calibration signal S _C to the sensing circuit.

The variable resistor is adjustable, thus changing the sensitivity of the sensor system. Preferably, the sense resistor has a range of operating resistance (the range has a fixed magnitude), and adjustments to the variable resistor change the position of the range of operating resistance of the sensing resistor, i.e. the lower end of the range of operating resistance point. This in turn affects the voltage output of the sensing circuit in the absence of pumping, which affects the sensitivity of the system. In a preferred embodiment, the variable resistor is adjusted such that the operating resistance range of the sense resistor has a low point at or just below zero. This provides the best sensitivity.

A self-adjusting offset circuit can be used to vary the sensitivity of the sensor system. An offset circuit can be formed by connecting the output of the microcontroller to the non-inverting input of a differential amplifier and connecting the output of the differential amplifier to the input of the microcontroller. The microcontroller can monitor the output of the differential amplifier, V _OUT , and provide a voltage to the non-inverting input until V _OUT =0.

The pulsed calibration signal S _C can be used to adjust the sensitivity of the sensor system. Preferably, the width of each pulse of the pulsed drive signal S ₁ is adjusted for each pulse of the calibration signal S _C . This adjustment is preferably arranged to vary the proportion of each pulse of the signal _S1 in which a change in the voltage output of the sensing circuit indicative of a puff can be detected. The pulsed calibration signal S _C may be set to operate at the first frequency or the second frequency whenever it has a pulse at the xth pulse of the pulsed drive signal S ₁ . x is any suitable value, such as 1000. Alternatively, the pulsed calibration signal S _C may be set to have a pulse whenever the pulsed drive signal S ₁ switches from the first frequency to the second frequency or at other suitable times. If the means for providing the pulsed drive signal S ₁ comprises a microcontroller, preferably the pulsed calibration signal _Sc is provided at the output of the microcontroller.

The sense resistor may be a silicon MEMS based resistor. In another embodiment, the sense resistor may form part of a silicon MEMS based sensor. The sensor may also include a reference resistor.

The sensing circuit may include a Wheatstone bridge having a first branch and a second branch, and wherein the voltage output is the difference between the voltage across the first branch and the voltage across the second branch.

According to a second aspect of the present invention there is provided an aerosol generating system for receiving an aerosol forming substrate, the system comprising an aerosol generating system for sensing fluid flow in an aerosol generating system indicative of draw according to the first aspect of the present invention flow sensor system.

The aerosol generating system may be an electrically heated aerosol generating system. The aerosol generating system may be a smoking system. Preferably, the system is portable. Preferably, the system includes a housing for receiving the aerosol-forming substrate and designed to be held by a user.

The aerosol-forming substrate may comprise a tobacco-containing material that includes volatile tobacco flavor compounds that are released from the substrate upon heating. The aerosol-forming substrate also includes an aerosol former. The aerosol forming substrate may be a fixed substrate, a liquid substrate, a gas substrate, or a combination of two or more of solid, liquid and gas.

If the aerosol-forming substrate is a liquid substrate, the aerosol generating system may include an aerosolization mechanism in contact with a source of the liquid substrate. The aerosolization mechanism may include at least one heating element for heating the substrate to form an aerosol; wherein the heating element may be activated when the aerosol generating system senses fluid flow indicative of a draw. Alternatively, the heating element may be separate from, but in communication with, the aerosolization mechanism. The at least one heating element may comprise a single heating element or more than one heating element. The heating element may take any suitable form to most efficiently heat the aerosol-forming substrate. The heating element preferably comprises a resistive material.

The aerosolization mechanism may include one or more electromechanical elements such as piezoelectric elements. The aerosolization mechanism may include elements using electrostatic, electromagnetic or pneumatic effects. The aerosol generating system may include a condensation chamber.

During operation, the substrate may be fully contained within the aerosol generating system. In this case, the user can draw on the nozzle of the aerosol generating system. Alternatively, the substrate may be partly contained within the aerosol generating system during operation. In this case, the substrate may form part of a separate item and the user may pump directly on the separate item.

The aerosol generating system may include a power source. The power source may be a lithium-ion battery or one of its variants, such as a lithium-ion polymer battery, or a nickel metal hydride battery, a nickel cadmium battery, a supercapacitor, or a fuel cell. In an alternative embodiment, the aerosol generating system may include a circuit rechargeable by an external charging portion and arranged to provide power for a predetermined number of puffs.

According to a third aspect of the present invention there is provided a method for actuating a flow sensor system for sensing fluid flow indicative of a draw in an aerosol generating system, the sensor system being arranged in a first mode and Operating in a second mode, in the first mode, no puff is expected or detected, and in the second mode, a puff is expected or detected, the method comprising the steps of: providing a sensing circuit for sensing A circuit powered pulsed drive signal S ₁ such that the sensing circuit is powered when the pulsed drive signal S ₁ is high and unpowered when the pulsed drive signal S ₁ is low, the sensing circuit comprising a sense resistor and a voltage output, the sense resistor being arranged to detect fluid flow based on a change in resistance of the sense resistor, the sense circuit being arranged such that the change in resistance of the sense resistor causes a change in voltage output; and at The sensor system is switched between a first and a second mode of operation, wherein the pulsed drive signal S ₁ has a first frequency f ₁ in the first mode and a second frequency greater than the first frequency f ₁ in the second mode. frequency f ₂ .

Driving the flow sensor system with a pulsed drive signal S ₁ means that the sensing circuit is not powered continuously, but only when S ₁ is high. This reduces power consumption significantly, while _fi and _f2 can be chosen for proper sensitivity.

In one embodiment, switching the sensor system between the first and second modes of operation comprises switching the sensor system from a first mode in which the pulsed drive signal S 1 has a first frequency f 1 to a mode in which the pulsed drive signal S ₁ has a first frequency f ₁ when a puff is detected. The drive signal S ₁ has a second mode with a second frequency f ₂ . The puff is detected by a change in the voltage output of the sensing circuit. Alternatively, or in addition, switching the sensor system between the first and second modes of operation includes switching the sensor system from the first mode in which the pulsed drive signal S ₁ has the first frequency f ₁ based on user habits when a puff is expected to occur. The mode is switched to a second mode in which the pulsed drive signal S ₁ has a second frequency f ₂ . The time at which a puff is expected to occur may be predicted based on user habits. For example, the sensor system may be switched from the first mode to the second mode at one or more of a predetermined time period after a previous puff and a predetermined time of day. For the user, the predetermined period of time may be an average time between puffs, and it may be adaptive such that it is continuously adjusted based on a moving average of the time between puffs. Alternatively, the predetermined period of time may have a fixed value. This is advantageous because the response time will be much shorter if the sensor system is operated in the second mode prior to puffing.

Preferably, the method comprises providing a pulsed drive signal S at _{a second} frequency f for a predetermined period of time after a change in the voltage output of the sensing circuit indicative of a puff has been detected, and at all other times at the first frequency f ₁ to provide the pulse driving signal S ₁ .

Preferably, the method further comprises the step of providing a signal _S2 to other components in the aerosol generating system, the signal _S2 being high when the voltage output of the sensing circuit indicates that a puff is detected, and the signal _S2 being high when sensing The voltage output of the circuit is low indicating no puff is detected. Signal _S2 may be used to activate: one or more of: an aerosolization mechanism, a nebuliser, a heating element and a puff indicator.

The method may also include the step of adjusting the sensitivity of the sensor system, including one or more of: periodically adjusting the resistance of a variable resistor in the sensing circuit; providing a self-adjusting offset circuit; The sensing circuit provides a pulsed calibration signal S _C .

The method may also include the step of delivering aerosol to the user based on a characteristic of the puff detected by the sensing circuit. Features described in relation to one aspect of the invention may also be applicable to another aspect of the invention.

The invention will be further described, by way of example only, with reference to the accompanying drawings, in which:

Figure 1 shows an exemplary embodiment of a sensor system according to the invention;

Figure 2a shows the signal GP2 of Figure 1;

Figure 2b shows the signal V _OUT of Figure 1 in the absence of pumping;

Figure 2c shows the signal V _OUT of Figure 1 when a puff is detected;

Figure 3 shows an alternative arrangement of the sensing circuit of Figure 1 in the form of a Wheatstone bridge;

Figure 4 shows how a relaxation set point can be established; and

FIG. 5 illustrates a method of operation of the sensor system of FIG. 1 .

A suitable sensor for use in the sensor system of the present invention may include a silicon substrate, a thin film of silicon nitride on the substrate, and two platinum heating elements on the thin film. The two heating elements are resistors, one acts as both an actuator and a sensor, and the other acts as a reference. Such sensors are advantageous because they provide fast sensor response. Of course, other suitable sensors may be used. During operation, there is a change in resistance of the sense resistor due to cooling by nearby fluid flow. This change in resistance is due to heat loss.

The sense resistor can be used at constant temperature, in which case the measurement requires increased heating power and provides an indication of fluid flow. Alternatively, the sensing sensor may be used at constant heating power, in which case the reduced temperature provides an indication of fluid flow. Alternatively, as will be described below with reference to Figures 1 and 3, a sense resistor may be used with a constant current, in which case a change in the balance of the sense circuit provides an indication of fluid flow.

FIG. 1 shows an exemplary embodiment of a sensor system according to the invention. The sensor system 101 of FIG. 1 comprises a sensing circuit 103, a predetermined value current source in the form of a current mirror 105, a differential amplifier 107, and a circuit for providing a pulsed drive signal S in the form of a microcontroller 109 and a drive transistor 111. ₁ signal generator.

The sensor system 101 of FIG. 1 includes a sensing circuit 103 . Sensing circuit 103 includes resistors R ₁ , R ₄ and variable resistor R _V in the left branch and resistors R ₂ , R ₃ and sensing resistor R _S in the right branch. The sense resistor R _S is the sense resistor of a sensor as described above or another suitable type of sensor. As will be described further below, _RV is an adjustable resistance and can be used to establish a relaxation set point (eg, when there is no gas flow in the system). Alternatively, a self-adjusting offset circuit can be used to establish the relaxation set point. In this embodiment, the output of the microcontroller is connected to the non-inverting input of a differential amplifier (not shown in FIG. 1 ), and the V _OUT of the differential amplifier can be connected to the input of the microcontroller. A microcontroller can be used to monitor the output of the differential amplifier, V _OUT , and inject a voltage on the non-inverting input of the differential amplifier until V _OUT =0.

等于右侧支路中的电阻的比

这导致V_DIFF＝V₂-V₁等于零。一旦R_S被流体流动冷却，则存在R_S的电阻变化，这导致右侧支路B的电压变化和V_DIFF的非零值。The measured voltage V _DIFF is a differential measurement (in this exemplary case, the difference between V ₂ in the right branch B and V ₁ in the left branch A). When the sensing circuit 103 is in a balanced state, the ratio of the resistances in the left branch

Equal to the ratio of the resistors in the right branch

This results in V _DIFF =V ₂ -V ₁ equal to zero. Once _R is cooled by the fluid flow, there is a change in the resistance of _R which results in a change in voltage in the right branch B and a non-zero value of _V.

For the sensing circuit 103 of FIG. 1, it can be easily shown that:

${\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; DIFF</span>}}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="V"\; filenames="HDA0000042982420000041.png"\; state="\{\{state\}\}">V</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; vs.</span>}<span\; class="notranslate">\; (</span>\frac{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; S</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="3"\; label="R"\; filenames="HDA0000042982420000021.png"\; state="\{\{state\}\}">R</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 3</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; -</span>\frac{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; v</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; 4</span>}}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; v</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>$

If R _v + R ₄ = R ₁ and R _s + R ₃ = R ₂ , then

${\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; DIFF</span>}}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="V"\; filenames="HDA0000042982420000041.png"\; state="\{\{state\}\}">V</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; vs.</span>}<span\; class="notranslate">\; (</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 0</span>}$

The differential measurement V _DIFF provides an indication of the fluid flow that caused the change in the resistance of R _S . Since V _DIFF is a differential measurement, very accurate measurements can be made even for small changes in fluid flow and thus resistance. This structure allows recording information such as puff volume and intensity. Note that V _DIFF is not linearly related to sense resistor R _S according to equation (1).

In the embodiment of FIG. 1 , the predetermined value current source takes the form of a current mirror 105, which includes two transistors T ₁ and T ₂ in a mirror structure, plus a resistor R _REF . The current _IM at _T2 must be equal to I _REF at _T1 (which is also the current through the sense circuit 103). And: V _S = R _REF I _REF + V _BE

therefore:

${\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; REF</span>}}<span\; class="notranslate">\; =</span>\frac{{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; BE</span>}}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; REF</span>}}}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; )</span>$

Non-linearities in the sensing circuit are compensated by the current mirror (see equations (1) and (2) above). This is advantageous because in systems where the nonlinearities are compensated in this way, the nonlinearities are found to be two times smaller than in systems compensated by voltage variations. Therefore, the current mirror 105 in the embodiment of FIG. 1 reduces the non-linearity of the system.

Current mirror 105 may have any suitable structure. A current mirror may be placed on the high voltage side of the sensing circuit 103 instead of between the sensing circuit and ground as shown in FIG. 1 . Any suitable transistor type may be used for _T1 and _T2 , including PNP transistors, NPN transistors and CMOS transistors. There may also be alternative settings for the current source. The sensor system must operate correctly within a reasonable temperature range, and the current mirror 105 compensates for any temperature changes. Other temperature compensated current sources are also available. If the external temperature changes, the output voltage V _DIFF of the sensing circuit will be affected, which may cause inaccurate operation or measurement. _T1 and _T2 should have the same electrical characteristics and be placed close together and in similar packages so as to minimize any temperature difference between them.

On the other hand, with reference to the particular setup of the current mirror 105, if there is a temperature difference between _T1 and _T2 , then since both transistors have the same potential (V _BE ) across their base-emitter junctions, V _BE keep constant. This means that if the two transistors are at different temperatures, the current through _T1 is different than the current through _T2 in order to maintain _VBE . On the other hand, if the external temperature changes to affect _T1 and _T2 equally, the current through both transistors changes equally, keeping _V constant.

The sensor system 101 also includes a differential amplifier 107 at the output of the sensing circuit 103 in order to amplify the output voltage V _{DIFF which} is typically only a few millivolts. In Fig. 1, an AD623 amplifier manufactured by Analog Devices, Massachusetts, USA is used. These amplifiers use less than 0.5mA and have gains as high as 1000. However, any suitable differential amplifier may be substituted. Amplifier 107 is connected to supply voltage _VS , and the gain of the amplifier is set by resistor _RG according to the following equation:

${\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; out</span>}}<span\; class="notranslate">\; =</span><span\; class="notranslate">\; \{</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; 100000</span>}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; G</span>}}}<span\; class="notranslate">\; \}</span>{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; DIFF</span>}}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 3</span>}<span\; class="notranslate">\; )</span>$

Therefore, for a gain of ≈1000, _RG is set to 100Ω.

Equation (3) applies only over a certain range of V _DIFF . Either side of this range, the amplifier will saturate. In one example, if V _DIFF =0V, then V _OUT =1.5V. If V _DIFF < -1.5mV, then V _OUT saturates at 0V. If V _DIFF >+1.5mV, then V _OUT saturates at 3V. In the range -1.5mV < V _DIFF < +1.5mV, equation (3) holds, that is, the relationship is linear with a gradient equal to the gain, which is about 1000 if R _G is set to 100Ω .

The sensor system 101 also includes a microcontroller 109 and a drive transistor 111 . In one embodiment, the microcontroller has an input GP0 and outputs GP2 and GP4. In FIG. 1 , the sensing circuit 103 and circuit mirror 105 are the highest power consumers. In order to reduce power consumption, the sensing circuit 103 and the current mirror 105 are not continuously powered, but driven by a pulsed drive signal S1 from the microcontroller 109 . The pulse current I _REF is supplied to the current mirror 105 and the sensing circuit 103 via the drive transistor 111 in accordance with the signal S ₁ on the output GP2 of the microcontroller 109 . The driving transistor 111 functions as a switch and is turned on when the GP2 signal is high. The width and frequency of the pulses are controlled by microcontroller 109 . In this embodiment, the output V _OUT is connected to the input GP0 of the microcontroller to digitize the differential amplifier output. The output GP0 is observed, and the width and frequency of the pulse signal at GP2 can be adjusted accordingly. In the embodiment of FIG. 1 , the microcontroller 109 is a CMOS Flash-based 8-bit microcontroller of the PIC12f675 series manufactured by Microchip Technology, Arizona, USA. The microcontroller has a power port, a ground port, and six input/output (I/O) ports GP0 to GP5, including four analog-to-digital conversion ports. It can run at 3V. Of course, any suitable microcontroller may be used.

Figure 2a shows one pulse of the square wave signal (signal S1) at the microcontroller output GP2. Figure 2b shows how the GP2 signal affects the signal at _VOUT in the absence of pumping. Figure 2a shows a graph of voltage versus time for GP2. Figure 2b shows a graph of voltage for V _OUT versus time. The graphs of Figures 2a and 2b are not drawn to scale. Each pulse of the signal at GP2 in Figure 2a is divided into three phases labeled f, g and h in Figure 2a. These stages are discussed below. The signal at V _OUT in Figure 2b is divided into five phases labeled a, b, c, d and e in Figure 2b.

In phase a, the GP2 signal is 0V. This is before the pulse. Therefore, no current is supplied to the sensing circuit 103 . The sense resistor R _S has no current flowing through it, so it is at ambient temperature. The output V _DIFF of the sensing circuit 103 is 0V, which produces an output V _OUT of 1.5V as discussed above.

In phase b, the GP2 signal is 3V. Current is now supplied to the sensing circuit 103, which means that the temperature of R _S starts to increase. The output V _DIFF of the sensing circuit 103 increases to more than 1.5mV, which means that the amplifier output V _OUT saturates at 3V.

In phase c, the temperature of R _S continues to rise and this starts to reduce the output of the sensing circuit 103 . V _DIFF falls below the saturation level of 1.5mV, allowing a linear response from the amplifier output, V _OUT . Therefore, V _OUT decreases linearly with V _DIFF as R _S temperature rises.

In phase d, the temperature of R _S has risen sufficiently that V _DIFF is less than -1.5mV and the amplifier output V _OUT saturates again, this time at 0V.

In phase e, the pulse on GP2 has ended, so the voltage of GP2 is at 0V again. Current is no longer supplied to the sensing circuit 103, which means the output V _DIFF is 0V, which results in an output V _OUT of 1.5V, as in phase a. R _S cools down before the next pulse.

In this system, pumping can be detected during phase c of V _OUT , ie during the linear response of the differential amplifier. In a conventional setup, the sensing circuit 103 is set up such that its equilibrium V _DIFF =0 is reached when the sensor heater resistance reaches a constant temperature at zero flow. In the case of constant current, this means supplying current to the sensor long enough for the sensor heater resistance to reach equilibrium temperature. This means high power consumption of the sensor. In this embodiment of the invention, power consumption is reduced by setting the pulses such that the heater resistance cannot or is just able to reach its equilibrium temperature.

Figure 2c shows the signal at V _OUT when a puff is detected. Figure 2c shows a graph of voltage for V _OUT versus time. Likewise, the graph of Figure 2c is not drawn to scale. When pumping is applied, the resulting fluid flow causes the V _OUT ramp (phase c) to shift to the right. The amount of displacement of the ramp is proportional to the flow velocity. As the ramp is shifted to the right, this ultimately provides a V _OUT signal of the same form as the GP2 drive signal shown in Figure 2a. This is shown in Figure 2c. The GP2 signal goes to zero before or simultaneously with the start of the phase c ramp. A puff is detected immediately before the end of the GP2 pulse. If the V _OUT signal is digitized (via GP0 ), a puff is considered detected if its value is above a threshold. Therefore, it is important that V _OUT is 0V without any fluid flow and before measurement.

Figure 3 shows an alternative arrangement of the sensing circuit 103 in the form of a quarter Wheatstone bridge structure 303 comprising a sensing resistor _RS . The four sides of the Wheatstone bridge include resistors R ₁ , R _V (in left leg A'), R ₂ and (R ₃ + _RS ) (in right leg B'), respectively. Likewise, R _V is an adjustable resistance and is used to establish the set point of the Wheatstone bridge. This bridge arrangement is advantageous because it allows detection of small changes in sensor resistance. Additionally, this setup reduces variations due to changes in ambient temperature.

Figure 4 shows how a variable resistor RV or a self-adjusting offset circuit can be used to establish a relaxation set point for the sensing circuit 103 or Wheatstone bridge 303 and adjust the sensitivity of the sensor system. As described with reference to stages b, c and d of Fig. 2b, the sensor resistance _RS rises to a value determined by the pulse width of the signal GP2 generated by the microcontroller 109 when powered on. R _V or a self-adjusting offset circuit can be used to determine at what voltage level this change in _R occurs, and this is illustrated in Figure 4.

The range of values that R _S can take is shown in FIG. 4 as range 401 in the case of temperature variations. The effect of the R _V adjustment and use of the self-adjusting offset circuit is to move the range 401 diagonally, as indicated by arrow 403 . The relaxation set point is the point at which the voltage change of _RS is set. The movement of the range 401 of R _S along the diagonal in Fig. 4 corresponds to the shift from left to right of the ramp of phase c of V _OUT in Fig. 2b. The best sensitivity is achieved when range 401 starts at or just below zero in Figure 4 (this corresponds to the V _OUT phase c ramp at or immediately before the end of the GP2 pulse in Figure 2b).

FIG. 5 illustrates one embodiment of a method of operation of the arrangement of FIG. 1 . The upper third of FIG. 5 shows a graph of voltage versus time for GP2 (signal S1 ). The middle third of FIG. 5 shows a graph of voltage versus time for V _OUT (corresponding to GPO). The lower third of FIG. 5 shows a graph of voltage versus time for the microcontroller output V _CTRL (corresponding to signal S ₂ on GP4 ). The graphs in Figure 5 are not drawn to scale. As previously discussed, to minimize power consumption, the sensing circuit 103 or Wheatstone bridge 303 and current mirror 105 are powered with the pulsed drive signal S1 at GP2. A square pulse of GP2 is shown in FIG. 2a. The left side of Figure 5 shows the signals operating in the first mode. The right side of Figure 5 shows the signals operating in the second mode.

The left side of Figure 5 shows the method of operation when no puff is detected and the signal is operating in the first mode. In this embodiment, the pulse frequency when the signal is operating in the first mode is 3 Hz, ie approximately one pulse every 330 ms. This frequency provides a good compromise between sensitivity and power consumption. In this embodiment, the pulse width of GP2 is 12.1 ms. Accordingly, voltage V _OUT has the form shown on the left side of FIG. 5 . Note that each pulse of V _OUT in the bottom half on the left side of Fig. 5 has the shape shown in Fig. 2b, but the pulse shape is only shown schematically in Fig. 5 . In the left side of Fig. 5, no puff is detected, so the pulse shape is similar to that shown in Fig. 2b instead of Fig. 2c.

The right side of Figure 5 shows the method of operation when a puff is detected and the signal is operating in the second mode. A puff is detected at time 501 . As can be seen in the right middle third of Figure 5, pumping is detected because the second half of the V _OUT pulse (bottom of the phase c ramp) has a higher value. This corresponds to a fluid flow that shifts the ramp in phase c to the right such that the ramp is cut off by the GP2 signal returning to 0V before reaching phase d. When a puff is detected at time 501, detection of input GP0 switches GP4 output signal _S2 from 0 to 1, causing V _CTRL to turn on, as shown in the lower right third of FIG. 5 . Detection of input GP0 also causes a change in the pulse frequency of GP2 and the system starts operating in the second mode. Of course, GP4 signal changes can also be used to control other circuits such as the aerosolization mechanism, nebulizer, heating element and puff indicator. Now, in this embodiment, the pulse frequency of GP2 in the second mode is 22 Hz, or about one pulse every 45 ms, as shown in the upper third of the right side of FIG. 5 . Note that the pulse width is still the same as in the first mode, ie 12.1 ms in this embodiment. Note that the lower portion of the V _OUT signal follows the dashed curve labeled 503 . This curve is a suction profile because the amount the slope of V _OUT is shifted to the right is proportional to the flow rate. As the lower portion of the V _OUT signal increases, the flow rate increases from zero to its maximum value, and as the lower portion of the V _OUT signal decreases from its maximum value to zero, the flow rate decreases from its maximum value to zero.

In this embodiment, the system has been properly calibrated; this can be seen from curve 503, which only approaches but does not exceed high values of V _OUT . This is equivalent to the extent of the R _S 401 starting at or just below zero and the V _OUT phase c ramp at or just before the end of the GP2 pulse in FIG. 4 . This calibration can be achieved by a change in _RV or offset circuit as discussed above with reference to FIG. 4 or by an alternative calibration method as will be discussed below.

At time 505, when no change is detected again at V _OUT , the output V _CTRL returns to 0V. After a puff is detected at time 501 , the GP2 pulse remains at the second frequency of 22 Hz for a predetermined period of time until time 507 when it returns to the first frequency of 3 Hz. This period of time 501 to 507 may be preset or based on user habits. For example, the time period may correspond to an average time period between two puffs.

Thus, during the first mode, when the GP2 pulse frequency is 3Hz, the time to the first puff is about 330ms in the worst case. If puffing is done during the second mode, the maximum response time is much faster when the GP2 pulse frequency is 22Hz, and in the worst case the time to puff is about 45ms.

The V _OUT signal indicative of the puff can be recorded and used to determine various data. For example, the average total time spent on puffs can be recorded from the V _OUT signal. This is time 501 to 505 in FIG. 5 . Additionally, the slope of curve 503 can be used to calculate the force or intensity with which the user takes an inhalation. Furthermore, the amount of puffs may be determined from puff distribution 503 over time 501-505. Additionally, the average time between puffs can be recorded from the V _OUT signal (but note that only one puff is shown in Figure 5 for simplicity).

This information can be fed into a microcontroller, and this allows a great deal of flexibility in operation. For example, based on the recorded time between puffs, the microcontroller can modify the time period during which the GP2 remains at high frequency (501 to 507) according to the user's habits. As another example, the microcontroller may automatically switch from low frequency GP2 pulses to high frequency GP2 pulses based on the user's habits at the time when the next puff is expected. This will reduce the response time, ie the time to puff. As another example, the force with which a user takes an inhalation can be recorded and used to manage aerosol delivery at eg the actuator, aerosolization mechanism or heating element to suit the user.

The method of operation shown in Figure 5 can be implemented with microcontroller software. First, the software powers up and initializes the microcontroller. Next, the software performs electronic stabilization. Once these processes have been completed, the microcontroller can be used to generate pulses on GP2 and read the response at _VOUT . If V _OUT is not greater than 0.1V, no puff is detected, in which case GP2 signal S ₁ is set to be at the first pulse frequency, in this case 3 Hz. The microcontroller continues to generate pulses at the first pulse frequency and reads the response at V _OUT until a puff is detected.

If V _OUT is greater than 0.1V, a puff is detected, in which case a countdown timer is started. This corresponds to time 501 in FIG. 5 . The microcontroller output V _CTRL to GP4 (S ₂ ) is set high and the GP2 signal is set at a second pulse frequency, in this case 22Hz. The microcontroller generates a pulse at the second frequency on GP2 and reads the response at _VOUT . If V _OUT is greater than 0.1V, a puff is still detected, in which case the GP2 pulse S ₁ is still issued at the second frequency and the microcontroller output V _CTRL to GP4 (S ₂ ) remains high.

If V _OUT is not greater than 0.1V, then the puff is no longer detected. This corresponds to time 505 in FIG. 5 . In this case, V _CTRL is set low. Then, if the countdown timer is non-zero, the time period during which the GP2 pulse should remain at high frequency has not expired, ie time 507 in FIG. 5 has not been reached. In this case, the GP2 pulse signal _S1 is kept at a high frequency.

If the countdown timer is zero, the time period during which the GP2 pulse should remain at high frequency has expired, ie time 507 in FIG. 5 has been reached. In this case, the GP2 pulse signal S ₁ returns to the first low frequency.

As discussed above, the sensitivity of the system can be set by adjusting _RV or by injecting a voltage into the non-inverting input of the difference amplifier until the output of the amplifier _VOUT is 0V. Another way is to use the calibration signal S _C . The calibration signal S may be generated periodically, eg every x pulses (eg 1000 pulses) of signal _S1 on GP2, or whenever the GP2 signal changes from the second mode (22 Hz) to the first mode (3 Hz) Pulse on _C. Referring again to Figure 2a, a calibration pulse is used to maintain a constant period of time for phase d (ie when V _OUT is 0V). If a calibration pulse is used, the pulse width of GP2 is no longer fixed but variable. As shown in Figure 2a, the GP2 pulse is divided into three phases f, g and h. During calibration, the GP2 signal is held high at 3V in a phase f of fixed duration (6 ms in one embodiment), regardless of the V _OUT signal. In phase g, the V _OUT signal is monitored and the GP2 signal remains high at 3V as long as V _OUT remains greater than 0V (also in phase b or c - see Figure 2b). Once the V _OUT signal reaches 0V (phase d - see Figure 2b), this time is recorded and the period of phase h of GP2 corresponding to phase d of V _OUT is set at a fixed duration (300 μs in one embodiment ). During calibration, in this embodiment, a puff is considered detected if V _OUT does not reach 0V after a total pulse time (f+g+h) of 14ms.

In normal operation mode, the total pulse width of GP2 is f+g+h. The time g recorded during calibration is now used for the calculation of the total pulse width. This method of calibrating the system to set the sensitivity is advantageous for the following reasons. First, the adjustable resistor R _V can be replaced by a fixed resistor. Second, auto-calibration occurs whenever the pulsed calibration signal S _C has a pulse. This means that there is no need to manually adjust any components in the system during manufacture or during maintenance as the system itself will automatically adjust to optimum sensitivity. The time window of from 6 ms to 14 ms chosen in this embodiment is large enough to allow for any changes in ambient temperature and the response of the various electronic components, but any suitable time window may be chosen.

Claims (14)

1. one kind is used for indicating the mobile flow sensor systems of fluid of aspirating at smog generation systems sensing, this sensing system is configured to inexpectancy therein or detects under first pattern of suction and wherein expect or detect under second pattern of suction and operate, and comprises:

Sensing circuit, it comprises sense resistor and voltage output, and described sense resistor is configured to come test fluid to flow based on changes in resistance, and described sensing circuit is configured such that the resistance variations of sense resistor causes the variation of voltage output; And

Signal generator, it is configured to provide pulse drive signal S to described sensing circuit ₁So that, make described sensing circuit at pulse drive signal S to described sensing circuit power supply ₁Be powered when high, and at pulse drive signal S ₁Be not powered when low, wherein, pulse drive signal S ₁Under first pattern, has first frequency f ₁, and under second pattern, have greater than first frequency f ₁Second frequency f ₂

2. flow sensor systems according to claim 1 also comprises being configured to provide the current source of predetermined value electric current by described sensing circuit, wherein, and pulse drive signal S ₁Be provided for this current source.

3. according to claim 1 or the described flow sensor systems of claim 2, also comprise the differential amplifier that is configured to the voltage output amplification of described sensing circuit.

4. flow sensor systems according to claim 3, wherein, differential amplifier output is output into ratio with the voltage of sensing circuit in the certain limit of the voltage output value of described sensing circuit, and the voltage output of sensing circuit less than or saturated during greater than this scope.

5. according to each described flow sensor systems in the aforementioned claim, also comprise the device of the sensitivity that is used to adjust described sensing system, the described device that is used for adjusting sensitivity comprises the one or more of the following:

Variable resistance in the described sensing circuit;

The self-adjusting off-centre circuit; And

Signal generator, it is used for providing pulse matching signal S to described sensing circuit _C

6. according to each described flow sensor systems in the aforementioned claim, wherein, described sensing circuit comprises the Wheatstone bridge with first branch road and second branch road, and wherein, and described voltage output is to stride the voltage of first branch road and stride poor between the voltage of second branch road.

7. smog generation systems that is used to admit smog to form matrix, this system comprise according in the aforementioned claim each be used for the flow sensor systems that the fluid in smog generation systems sensing indication suction flows.

8. smog generation systems according to claim 7 also comprises:

At least one heating element heater, it is used for described matrix is heated to form smog;

Wherein, described flow sensor systems is configured to activate heating element heater when the fluid of indicating suction flows when described flow sensor systems senses.

9. one kind is used for driving the method that is used in the mobile flow sensor systems of the fluid of smog generation systems sensing indication suction, this sensing system is configured to inexpectancy therein or detects under first pattern of suction and wherein expect or detect under second pattern of suction and operate, and said method comprising the steps of:

Provide pulse drive signal S to sensing circuit ₁So that, make described sensing circuit at pulse drive signal S to described sensing circuit power supply ₁Be powered at pulse drive signal S when high ₁Be not powered when low, described sensing circuit comprises sense resistor and voltage output, described sense resistor is configured to come test fluid to flow based on the resistance variations of described sense resistor, and described sensing circuit is configured such that the resistance variations of described sense resistor causes the variation of voltage output; And

Between first and second operator schemes, switch described sensing system, wherein, pulse drive signal S ₁Under first pattern, has first frequency f ₁, and under second pattern, have greater than first frequency f ₁Second frequency f ₂

10. method according to claim 9, wherein, the step of switching sensing system between first and second operator schemes is included in when detecting suction sensing system from pulse drive signal S wherein ₁Has first frequency f ₁First pattern switch to wherein pulse drive signal S ₁Has second frequency f ₂Second pattern.

11. according to claim 9 or the described method of claim 10, wherein, the step of switching sensing system between first and second operator schemes comprise based on user's custom when suction takes place in expection with sensing system from pulse drive signal S wherein ₁Has first frequency f ₁First pattern switch to wherein pulse drive signal S ₁Has second frequency f ₂Second pattern.

12., comprise that also other assembly in the smog generation systems provides signal S according to each the described method in the claim 9 to 11 ₂Step, signal S ₂When the voltage output indication of sensing circuit detects suction is high, and signal S ₂When the voltage output indication of sensing circuit does not detect suction is low.

13. according to each the described method in the claim 9 to 12, also comprise the step of the sensitivity of adjusting described sensing system, comprise one or more in the following:

Periodically adjust the resistance of the variable resistance in the described sensing circuit;

The self-adjusting off-centre circuit is provided; And

Provide pulse matching signal S to described sensing circuit _C

14., also comprise according to characteristic and transmit smog to the user by the detected suction of described sensing circuit according to each the described method in the claim 9 to 13.

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